Current control for a boost converter with dual anti-wound inductor

ABSTRACT

A system may include a power converter comprising at least one stage having a dual anti-wound inductor having a first winding and a second winding constructed such that its windings generate opposing magnetic fields in its magnetic core and constructed such that a coupling coefficient between the first winding and the second winding is less than approximately 0.95 and a current control subsystem for controlling an electrical current through the dual anti-wound inductor, the current control subsystem configured to minimize a magnitude of a magnetizing electrical current of the dual anti-wound inductor to prevent core saturation of the dual anti-wound inductor and regulate an amount of output electrical current delivered by the power converter to the load in accordance with a reference input signal.

RELATED APPLICATION

The present disclosure claims priority as a continuation-in-partapplication to U.S. Provisional Patent Application Ser. No. 16/709,036,filed Dec. 10, 2019, which in turn claims priority to U.S. ProvisionalPatent Application Ser. No. 62/783,547, filed Dec. 21, 2018, both ofwhich are incorporated by reference herein in their entireties.

FIELD OF DISCLOSURE

The present disclosure relates in general to circuits for audio devices,piezoelectric devices, haptic-feedback devices, and/or other devices,including without limitation personal audio devices such as wirelesstelephones and media players, and more specifically, to an augmentedmulti-stage boost converter that may be used in such devices.

BACKGROUND

Personal audio devices, including wireless telephones, such asmobile/cellular telephones, cordless telephones, mp3 players, and otherconsumer audio devices, are in widespread use. Such personal audiodevices may include circuitry for driving a pair of headphones, one ormore speakers, a piezoelectric transducer, a haptic feedback transducer,and/or other transducer. Such circuitry often includes a driverincluding a power amplifier for driving a transducer output signal tothe transducer. Oftentimes, a power converter may be used to provide asupply voltage to a power amplifier in order to amplify a signal drivento speakers, headphones, piezoelectric transducers, haptic feedbacktransducers, or other transducers. A switching power converter is a typeof electronic circuit that converts a source of power from one directcurrent (DC) voltage level to another DC voltage level. Examples of suchswitching DC-DC converters include but are not limited to a boostconverter, a buck converter, a buck-boost converter, an invertingbuck-boost converter, and other types of switching DC-DC converters.Thus, using a power converter, a DC voltage such as that provided by abattery may be converted to another DC voltage used to power the poweramplifier.

Battery-powered systems may use a boost converter to generate a powersupply for an audio amplifier that is greater than a voltage of thebattery. For example, a motivation for using a boost converter in abattery-powered transducer is to generate a greater signal swing at theoutput of a transducer amplifier than could be achieved by powering theamplifier directly from the battery.

Traditionally, while a boost converter and an amplifier powered from theboost converter are often manufactured on the same integrated circuit,boost converters often require a boost inductor external to theintegrated circuit, which requires significant space. However, recentadvances in manufacturing have enabled the integration of inductors witha magnetic core into an integrated circuit die. Advantages of anintegrated inductor may include smaller total circuit area, significantreduction in height in a direction perpendicular to a surface of theintegrated circuit, lower electromagnetic interference emissions, andless variation of inductor physical properties.

Despite the advances in inductor manufacturing, designing a boostconverter with an integrated inductor may be challenging. External boostconverter inductors for audio applications generally have inductancesbetween 1 μH and 2 μH and saturate at between 2.5 A and 4 A of current.However, a typical integrated inductor may have an inductance in therange of tens to hundreds of nanohenries with a current saturation limitat or less than 1 A. A typical boost converter for audio may supply 12Vinto a 101V load from a 4V battery supply. Thus, even assuming 100%efficiency, a standard boost converter design may draw 2.5 A inputcurrent, which is well beyond the saturation point of the integratedinductor. A multi-phase converter could be used to distribute thecurrent to multiple inductors, but the small inductance causes a largecurrent ripple that may still exceed the saturation constraint.

To use an integrated inductor, the design of a power converter mustovercome the limitations of its low inductance and low saturationcurrent. One solution to this problem is to use a multi-wound inductorwith a modified boost converter architecture.

A multi-wound inductor may be used to weaken the magnetic field in thecore and prevent early saturation. FIG. 1A depicts a multi-woundinductor 100 with two coils 102 a and 102 b wrapped around a commonmagnetic core 104. FIG. 1B depicts a cross-sectional side view ofinductor 100 depicting current flow in each of coils 102 a and 102 b,with “·” depicting a current I₁ flowing out of the page in a directionperpendicular to the plane of the page and with “X” depicting a currentI₂ flowing into the page in a direction perpendicular to the plane ofthe page. Coils 102 a and 102 b may be wound in opposite directions suchthat positive current generates opposite fields in each coil. Therefore,a total magnetic flux φ_(M) through magnetic core 104 may equal thedifference between the magnetic flux φ_(M1) from coil 102 a and themagnetic flux φ_(M2) from coil 102 b. Magnetic fluxes φ_(M1), φ_(M2), incoil 102 a, 102 b may be proportional to currents I₁ and I₂,respectively, in such coil 102 a, 102 b.

Inductor 100 may saturate when the magnetic field in magnetic core 104exceeds a threshold, B_(sat). The magnetic field may be proportional tothe total magnetic flux φ_(M) in magnetic core 104, which may thereforebe proportional to the difference in currents (e.g., I₁-I₂). As aresult, a saturation constraint for inductor 100 may be given as:

I _(diff) ^(sat) ≥|I ₁ −I ₂|  (1)

where I_(diff) ^(sat) is a difference between current I₁ and current I₂that saturates inductor 100 and may typically be around 0.5 A-1.0 A foran integrated inductor. Equation (1) above may only be valid for low tomoderate levels of current. FIG. 1C illustrates a saturation profile ofcurrent I₂ versus current I₁. Dashed lines depict saturation boundaries108 from equation (1) whereas the hatched region depicts the truesaturation region 110 defined by the boundary ABCDE. For low currents,the unsaturated region 112 is a strip along the main diagonal asdescribed by equation (1). However, at larger currents the unsaturatedregion 112 shrinks in width until, at very large currents, inductor 100is always saturated.

This effect may occur because the field cancellation between coils 102 aand 102 b may not be perfect, especially at their respective ends. Also,some inductor designs may use extra turns of one of coils 102 a, 102 bto control a coupling coefficient which may further reduce the fieldcancellation. As a result, inductor 100 may saturate even though thecurrent difference |I₁−I₂| is within its limits. Thus, the condition ofequation (1) may represent a necessary (but not a sufficient) conditionfor saturation. Instead, a sufficient condition for inductor 100 to beunsaturated is that currents I₁ and I₂ must lie in unsaturated region112 defined by points ABCDE.

A multi-wound inductor may extend the range of winding currents that maybe used before the device is saturated. For example, if current I₂ iszero, current I₁ may only extend to point E in FIG. 1C and remainunsaturated. However, with a properly chosen value for current I₂, therange of current I₁ can be extended to point D or even point C andremain unsaturated due to the field cancellation of currents I₁ and I₂.This range extension can be used to help with the saturation problem ofintegrated boost inductors. However, the boost architecture must also bedesigned to take advantage of the benefits of a multi-wound inductor.

FIG. 2 depicts one example of a single-stage boost converter 200 thatmay be used with a multi-wound inductor 100 and having a load 202.Single-stage boost converter 200 may use capacitor 204 to stabilize itsoutput voltage V_(out). A battery 206 may supply single-stage boostconverter 200 with an input voltage V_(in). Single-stage boost converter200 may comprise a plurality of switches 210, 212, 214, and 216, eachswitch having a gate G to receive a control signal to control theconductivity of such switch (e.g., to selectively open and close suchswitch). Such control signals may comprise pulse-width modulationcontrol signals labeled P₁ and P₂ in FIG. 2, along with theirrespectively logical complements, signals labeled P₁ and P₂ in FIG. 2.Switches 210 and 212 may toggle top coil 102 a of inductor 100 between acharging state in which coil 102 a is coupled between battery 206 andground and a transfer state wherein coil 102 a is coupled between powersupply 206 and load 202. Likewise, switches 214 and 216 may togglebottom coil 102 b of inductor 100 between a charging state in which coil102 b is coupled between battery 206 and ground and a transfer statewherein coil 102 b is coupled between power supply 206 and load 202. Theboost voltage ratio, V_(out)/V_(in), may be related to the pulse-widthmodulation duty cycle D of control signals P₁ and P₂ with an equationthat is very similar to that of a standard boost converter:

$\begin{matrix}{\frac{V_{out}}{V_{in}} = \frac{1}{1 - D}} & (2)\end{matrix}$

assuming no inductor or switching losses.

Single-stage boost converter 200 depicted in FIG. 2 may not prevent themulti-wound inductor from saturating at realistic boost voltages andoutput powers. For example, FIG. 3A depicts a circuit simulation ofcurrents I₁ and I₂ for single-stage boost converter 200 over onepulse-width modulation cycle, with an output voltage V_(out) of 12 V, anoutput power of 10 W, and an input voltage V_(in) of 4 V, which mayrepresent standard nominal operation conditions for a boost converter inan audio application. The simulation results as depicted in FIG. 3A alsomodel resistive losses in switches 210, 212, 214, and 216 and inductor100. FIG. 3B depicts current difference I₁−I₂ and saturation levelI_(diff) ^(sat) for inductor 100. FIG. 3C depicts currents I₂ versus I₁on a plot along with the saturation boundary I^(sat) also plotted inFIG. 3C, showing that although current difference I₁−I₂ remained belowsaturation level I_(diff) ^(sat) in FIG. 3B, their individual amplitudesexceeded saturation boundary I^(sat) in FIG. 3C. Accordingly,single-stage boost converter 200 may not be useful for a desiredapplication.

FIG. 4 depicts one example of a two-stage boost converter 400 that maybe used with multi-wound inductor 100 and having a load 202. Each stage401 a, 401 b of two-stage boost converter 400 may be identical tosingle-stage boost converter 200 shown in FIG. 2, and stages 401 a, 401b may be coupled in series. One disadvantage to two-stage boostconverter 400 is that it requires two capacitors, 204 and 205, tostabilize the output of each stage 401 compared to the single capacitor204 required for single-stage converter 200. Both capacitors 204 and 205may be large and may contribute significantly to the total circuit area.

In the architecture of two-stage boost converter 400, the boosted outputof first stage 401 a supplies the input voltage to second stage 401 b.Therefore, the total boost ratio of both stages 401 is the product ofthe boost ratio of each stage 401 a, 401 b. Because both stages 401 a,401 b may operate with identical duty cycles, the total boost ratio oftwo-stage boost converter 400 may be given as:

$\begin{matrix}{\frac{V_{out}}{V_{in}} = \left( \frac{1}{1 - D} \right)^{2}} & (3)\end{matrix}$

assuming no inductor or switching losses. Comparing equation (3) withequation (2), two-stage boost converter 400 may require a lower dutycycle than single-stage boost converter 200 to achieve the same boostvoltage ratio. For example, to boost from 4V to 12V, single-stage boostconverter 200 may require a duty cycle of 0.67 versus 0.42 for thetwo-stage boost converter 400. A lower duty cycle may decrease themagnitude of the current ripple, which should help prevent saturation.

FIG. 5A depicts a circuit simulation of currents I_(1-STAGE1),I_(2-STAGE1), I_(1-STAGE2), and I_(2-STAGE2) for two-stage boostconverter 400 over one pulse-width modulation cycle. A comparison ofFIG. 5A with FIG. 3A shows that coil current ripple amplitude may besignificantly reduced. FIG. 5B depicts current differenceI_(1-STAGE1)−I_(2-STAGE1), current difference I_(1-STAGE2)−I_(2-STAGE2),and saturation level I_(diff) ^(sat) for inductors 100. FIG. 5C depictscurrents I_(2-STAGE1) versus I_(1-STAGE1) and currents I_(2-STAGE2)versus I_(1-STAGE2) on a plot along with the saturation boundary I^(sat)also plotted in FIG. 5C. In comparing FIGS. 5B and 5C to FIGS. 3A and3B, respectively, the over-saturation issues of single-stage boostconverter 200 are shown to be greatly improved. The currents of inductor100 of first stage 401 a may now be within saturation limits. However,the currents of inductor 100 of second stage 401 b may still exceedsaturation limits. The problem with two-stage boost converter 400 may bethat even though the duty cycle is smaller, second stage 401 b issourced from the output of first stage 401 a, which is at a highervoltage than voltage V_(in) of battery 206. Therefore, inductor 100 ofsecond stage 401 b may experience a larger voltage drop when controlsignal P₁ is asserted, and that may cause the large current differenceI_(1-STAGE2)−I_(2-STAGE2) shown in FIG. 5B.

Thus, neither single-stage boost converter 200 nor two-stage boostconverter 400 may satisfy the saturation constraints of inductor 100 fordesired applications.

By including discussion in this Background section, Applicant is makingno admission that any of the content of this Background section is priorart that may be used to support a prior-art based rejection of therecited claims.

SUMMARY

In accordance with the teachings of the present disclosure, one or moredisadvantages and problems associated with existing inductor-based powerconverters may be reduced or eliminated.

In accordance with embodiments of the present disclosure, a system mayinclude a power converter comprising at least one stage having a dualanti-wound inductor having a first winding and a second windingconstructed such that its windings generate opposing magnetic fields inits magnetic core and constructed such that a coupling coefficientbetween the first winding and the second winding is less thanapproximately 0.95 and a current control subsystem for controlling anelectrical current through the dual anti-wound inductor, the currentcontrol subsystem configured to minimize a magnitude of a magnetizingelectrical current of the dual anti-wound inductor to prevent coresaturation of the dual anti-wound inductor and regulate an amount ofoutput electrical current delivered by the power converter to the loadin accordance with a reference input signal.

In accordance with these and other embodiments of the presentdisclosure, a method may be provided for controlling an electricalcurrent through a dual anti-wound inductor integral to a powerconverter, the dual anti-wound inductor having a first winding and asecond winding constructed such that its windings generate opposingmagnetic fields in its magnetic core and constructed such that acoupling coefficient between the first winding and the second winding isless than approximately 0.95. The method may include minimizing amagnitude of a magnetizing electrical current of the dual anti-woundinductor to prevent core saturation of the dual anti-wound inductor andregulating an amount of output electrical current delivered by the powerconverter to the load in accordance with a reference input signal.

In accordance with these and other embodiments of the presentdisclosure, a system may be provided for controlling an electricalcurrent through a dual anti-wound inductor integral to a powerconverter, the dual anti-wound inductor having a first winding and asecond winding constructed such that its windings generate opposingmagnetic fields in its magnetic core and constructed such that acoupling coefficient between the first winding and the second winding isless than approximately 0.95. The system may include an input forreceiving a reference input signal and a current control subsystemconfigured to minimize a magnitude of a magnetizing electrical currentof the dual anti-wound inductor to prevent core saturation of the dualanti-wound inductor and regulate an amount of output electrical currentdelivered by the power converter to the load in accordance with thereference input signal.

Technical advantages of the present disclosure may be readily apparentto one skilled in the art from the figures, description and claimsincluded herein. The objects and advantages of the embodiments will berealized and achieved at least by the elements, features, andcombinations particularly pointed out in the claims.

It is to be understood that both the foregoing general description andthe following detailed description are examples and explanatory and arenot restrictive of the claims set forth in this disclosure.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete understanding of the present embodiments and advantagesthereof may be acquired by referring to the following description takenin conjunction with the accompanying drawings, in which like referencenumbers indicate like features, and wherein:

FIGS. 1A and 1B depict a multi-wound integrated inductor, in accordancewith embodiments of the present disclosure;

FIG. 1C illustrates a saturation profile of currents within themulti-wound integrated inductor shown in FIGS. 1A and 1B, in accordancewith embodiments of the present disclosure;

FIG. 2 illustrates a single-stage boost converter using a multi-woundintegrated inductor, in accordance with embodiments of the presentdisclosure;

FIG. 3A depicts a circuit simulation of currents for the multi-woundintegrated inductor of the single-stage boost converter shown in FIG. 2over one pulse-width modulation cycle, in accordance with embodiments ofthe present disclosure;

FIG. 3B depicts a circuit simulation of a current difference and acurrent saturation level for the multi-wound integrated inductor of thesingle-stage boost converter shown in FIG. 2, in accordance withembodiments of the present disclosure;

FIG. 3C illustrates a saturation profile of currents within themulti-wound integrated inductor of the single-stage boost convertershown in FIG. 2, in accordance with embodiments of the presentdisclosure;

FIG. 4 illustrates a two-stage boost converter with each stage using amulti-wound integrated inductor, in accordance with embodiments of thepresent disclosure;

FIG. 5A depicts a circuit simulation of currents for the multi-woundintegrated inductors of the two-stage boost converter shown in FIG. 4over one pulse-width modulation cycle, in accordance with embodiments ofthe present disclosure;

FIG. 5B depicts a circuit simulation of a current difference and acurrent saturation level for the multi-wound integrated inductors of thetwo-stage boost converter shown in FIG. 4, in accordance withembodiments of the present disclosure;

FIG. 5C illustrates a saturation profile of currents within themulti-wound integrated inductors of the two-stage boost converter shownin FIG. 4, in accordance with embodiments of the present disclosure;

FIG. 6 illustrates selected components of an example personal mobiledevice, in accordance with embodiments of the present disclosure;

FIG. 7 illustrates a block diagram of selected components of an exampleintegrated circuit of a personal mobile device for driving a transducer,in accordance with embodiments of the present disclosure;

FIG. 8 illustrates a block and circuit diagram of selected components ofan example switched mode amplifier, in accordance with embodiments ofthe present disclosure;

FIG. 9 illustrates selected components of an augmented two-stage boostconverter with each stage using a multi-wound integrated inductor, inaccordance with embodiments of the present disclosure;

FIGS. 10A and 10B depict equivalent circuit diagrams showingconnectivity of selected components of the augmented two-stage boostconverter of FIG. 9 based on the values of switch control signals forthe augmented two-stage boost converter, in accordance with embodimentsof the present disclosure;

FIGS. 11A-11C depict a circuit simulation of currents for themulti-wound integrated inductors of the augmented two-stage boostconverter shown in FIG. 9 over one pulse-width modulation cycle, inaccordance with embodiments of the present disclosure;

FIG. 12A depicts an example model for modeling effects of disturbance ingeneration of pulse-width modulation control signals, in accordance withembodiments of the present disclosure;

FIG. 12B depicts an ideal pulse-width modulated control signal and theideal pulse-width modulated control signal affected by a disturbance, inaccordance with embodiments of the present disclosure;

FIGS. 13A and 13B depict a simulation of an example step disturbance inthe generation of pulse-width modulated control signals using thedisturbance model of FIG. 12A, in accordance with embodiments of thepresent disclosure;

FIG. 14A depicts magnetizing currents of inductors resulting from theresponse of the augmented two-stage boost converter shown in FIG. 9 tothe step disturbance depicted in FIGS. 13A and 13B along with amagnetizing current saturation limit, in accordance with embodiments ofthe present disclosure;

FIG. 14B illustrates a saturation profile of currents within themulti-wound integrated inductors of the augmented two-stage boostconverter shown in FIG. 9 during a cycle of operation prior to adisturbance in generation of a pulse-width modulated control signal, inaccordance with embodiments of the present disclosure;

FIG. 14C illustrates a saturation profile of currents within themulti-wound integrated inductors of the augmented two-stage boostconverter shown in FIG. 9 during a cycle of operation after adisturbance in generation of a pulse-width modulated control signal, inaccordance with embodiments of the present disclosure;

FIG. 15 illustrates an example voltage control loop that may be used inconnection with a boost converter, in accordance with embodiments of thepresent disclosure;

FIG. 16A illustrates selected components of an example current controlscheme that may be used in connection with a boost converter, inaccordance with embodiments of the present disclosure;

FIG. 16B illustrates selected components of an example control subsystemthat may be used in connection with a boost converter and the currentcontrol scheme depicted in FIG. 16A, in accordance with embodiments ofthe present disclosure;

FIG. 17A illustrates selected components, including a cycle averagecalculator, for performing calculation of cycle averages of inductorcoil currents, in accordance with embodiments of the present disclosure;

FIG. 17B illustrates selected components of a system for using atriangle carrier signal of a triangle modulator to generate pulse-widthmodulated control signals and using such triangle carrier signal totrigger a midpoint sampler configured to sample current values of themidpoint of the “ON” time of a pulse-width modulated control signals, inaccordance with embodiments of the present disclosure;

FIG. 17C illustrates a selected portion of a boost converter andselected components of a midpoint sampler, in accordance withembodiments of the present disclosure;

FIGS. 18A and 18B depict a simulation of an example step disturbance inthe generation of pulse-width modulated control signals, in accordancewith embodiments of the present disclosure;

FIG. 18C depicts example waveforms for cycle average currents ofinductor currents in response to the step disturbance of FIG. 18A, inaccordance with embodiments of the present disclosure;

FIGS. 19A-19C depict various waveforms for a three-cycle simulation ofcoordination of a midpoint sampler with a triangle modulator and thesampling process of FIG. 17C, in accordance with embodiments of thepresent disclosure;

FIGS. 20A and 20B depict waveforms showing an effect of offset caused byresistive losses and imbalance between boost converter stages, inaccordance with embodiments of the present disclosure;

FIGS. 21A and 21B illustrate selected components, including a forwardtransform block and a reverse transform block, for coordinatetransformation, in accordance with embodiments of the presentdisclosure;

FIG. 21C illustrates a saturation profile of currents within amulti-wound integrated inductor depicting transformed coordinate axes,in accordance with embodiments of the present disclosure;

FIGS. 22A-22D depict a simulation of an example step disturbance in thegeneration of pulse-width modulated control signals and an applicationof a forward transform to inductor coil currents, in accordance withembodiments of the present disclosure;

FIGS. 23A-23C depict selected components of an observer for use in acontrol subsystem for a boost converter, in accordance with embodimentsof the present disclosure;

FIGS. 24A-24D depict simulated actual values of pulse-width modulatedcontrol signal disturbances and coil currents and estimated versions ofsuch parameters as estimated by the Kalman filter shown in FIG. 23B, inaccordance with embodiments of the present disclosure;

FIG. 25A illustrates selected components of a control subsystem for aboost converter implementing two independent control loops formagnetizing and battery modes of the control subsystem, in accordancewith embodiments of the present disclosure; and

FIG. 25B depicts an example implementation of a state-space model for acontrol block of the control subsystem shown in FIG. 25A, in accordancewith embodiments of the present disclosure.

DETAILED DESCRIPTION

FIG. 6 illustrates an example personal mobile device 1, in accordancewith embodiments of the present disclosure. FIG. 6 depicts personalmobile device 1 having a speaker 7. Speaker 7 is merely an example, andit is understood that personal mobile device 1 may be used in connectionwith a variety of transducers including magnetic coil loudspeakers,piezo speakers, haptic feedback transducers, and others. In addition oralternatively, personal mobile device 1 may be coupled to a headset 3 inthe form of a pair of earbud speakers 8A and 8B. Headset 3 depicted inFIG. 6 is merely an example, and it is understood that personal mobiledevice 1 may be used in connection with a variety of audio transducers,including without limitation, headphones, earbuds, in-ear earphones, andexternal speakers. A plug 4 may provide for connection of headset 3 toan electrical terminal of personal mobile device 1. Personal mobiledevice 1 may provide a display to a user and receive user input using atouch screen 2, or alternatively, a standard liquid crystal display(LCD) may be combined with various buttons, sliders, and/or dialsdisposed on the face and/or sides of personal mobile device 1. As alsoshown in FIG. 6, personal mobile device 1 may include an integratedcircuit (IC) 9 for generating an analog signal for transmission tospeaker 7, headset 3, and/or another transducer.

FIG. 7 illustrates a block diagram of selected components of an exampleIC 9 of a personal mobile device for driving a transducer, in accordancewith embodiments of the present disclosure. As shown in FIG. 7, amicrocontroller core 18 may supply a digital input signal DIG_IN to adigital-to-analog converter (DAC) 14, which may convert the digitalinput signal to an analog input signal V_(IN). DAC 14 may supply analogsignal V_(I) to an amplifier 16 which may amplify or attenuate analoginput signal V_(I) to provide a differential audio output signal V_(O),which may operate a speaker, a headphone transducer, a piezoelectrictransducer, a haptic feedback transducer, a line level signal output,and/or other suitable output. In some embodiments, DAC 14 may be anintegral component of amplifier 16. A power supply 10 may provide thepower supply rail inputs of amplifier 16. In some embodiments, powersupply 10 may comprise a switched-mode power converter, as described ingreater detail below. Although FIGS. 6 and 7 contemplate that IC 9resides in a personal mobile device, systems and methods describedherein may also be applied to electrical and electronic systems anddevices other than a personal mobile device, including transducersystems for use in a computing device larger than a personal mobiledevice, an automobile, a building, or other structure.

FIG. 8 illustrates a block and circuit diagram of selected components ofan example switched mode amplifier 20, in accordance with embodiments ofthe present disclosure. In some embodiments, switched mode amplifier 20may implement all or a portion of amplifier 16 described with respect toFIG. 7. As shown in FIG. 8, switched mode amplifier 20 may comprise aloop filter 22, a controller 24, and a power converter 26.

Loop filter 22 may comprise any system, device, or apparatus configuredto receive an input signal (e.g., audio input signal V_(IN) or aderivative thereof) and a feedback signal (e.g., audio output signalV_(O), a derivative thereof, or other signal indicative of audio outputsignal V_(O)) and based on such input signal and feedback signal,generate a controller input signal to be communicated to controller 24.In some embodiments, such controller input signal may comprise a signalindicative of an integrated error between the input signal and thefeedback signal. In other embodiments, such controller input signal maycomprise a signal indicative of a target current signal to be driven asan output current I_(OUT) or a target voltage signal to be driven as anoutput voltage V_(O) to a load coupled to the output terminals of secondcontrol loop 28.

Controller 24 may comprise any system, device, or apparatus configuredto, based on an input signal (e.g., input signal INPUT), output signalV_(O), and/or other characteristics of switched mode amplifier 20,control switching of switches integral to power converter 26 in order totransfer electrical energy from a power supply V_(SUPPLY) to the load ofswitched-mode amplifier 20 in accordance with the input signal.

Power converter 26 may comprise any system, device, or apparatusconfigured to receive at its input a voltage V_(SUPPLY) (e.g., providedby power supply 10), and generate at its output output voltage V_(O). Insome embodiments, voltage V_(SUPPLY) may be received via input terminalsof power converter 26 including a positive input terminal and a negativeinput terminal which may be coupled to a ground voltage. As described ingreater detail in this disclosure (including, without limitation, inreference to FIGS. 9-14, below), power converter 26 may comprise a powerinductor and a plurality of switches that are controlled by controlsignals received from controller 24 in order to convert voltageV_(SUPPLY) to voltage V_(O), such that audio output signal V_(O) is afunction of the input signal to loop filter 22.

FIG. 9 depicts selected components of an example augmented two-stageboost converter 900 that may be used with multi-wound inductors 100 andhaving a load 202, in accordance with embodiments of the presentdisclosure. In some embodiments, augmented two-stage boost converter 900may be used to implement all or a portion of power supply 10 depicted inFIG. 7. In these and other embodiments, augmented two-stage boostconverter 900 may be used to implement all or a portion of powerconverter 26 depicted in FIG. 8. Augmented two-stage boost converter 900shown in FIG. 9 may be similar in many respects to two-stage boostconverter 400 depicted in FIG. 4, and thus, only differences betweenaugmented two-stage boost converter 900 and two-stage boost converter400 may be discussed below. In particular, while first stage 901 a ofaugmented two-stage boost converter 900 may be similar to first stage401 a of two-stage boost converter 400, augmented second stage 901 b ofaugmented two-stage boost converter 900, as compared to second stage 401b of two-stage boost converter 400, may include additional switches 910,912, 914, and 916 and capacitor 905 (in lieu of capacitor 405) arrangedas shown in FIG. 9 and controlled by control signals P₁, P₂, P₁ , and P₂as shown in FIG. 9. As in two-stage boost converter 400, inductors 100a, 100 b of each of stages 901 a and 901 b are dual, anti-woundinductors comprising a plurality of coils including coils 102 a and 102b and wound in such a manner that a magnetic field in a core 104produced by coils 102 a and 102 b cancels when currents through coils102 a and 102 b are positive.

FIGS. 10A and 10B depict equivalent circuit diagrams showingconnectivity of selected components of augmented two-stage boostconverter 900 based on the values of switch control signals foraugmented two-stage boost converter 900, in accordance with embodimentsof the present disclosure. In particular, FIG. 10A depicts connectivityof top coils 102 a of each of inductors 100 a and 100 b when controlsignal P₁ is asserted (and control signal P₁ is deasserted) and FIG. 10Bdepicts connectivity of top coils 102 a of each of inductors 100 a and100 b when control signal P₁ is deasserted (and control signal

$\overset{\_}{P_{1}}$

is asserted). For purposes of clarity of exposition, FIGS. 10A and 10Bneglect all resistive switch losses.

As seen in FIG. 10A, when control signal P₁ is asserted (and controlsignal P₁ is deasserted), top coils 102 a of inductors 100 a and 100 bare in parallel to the power supply of battery 206 and ground. Theconfiguration shown in FIG. 10A is a charging phase of augmentedtwo-stage boost converter 900 in which energy is stored in top coils 102a. As seen in FIG. 10B, when control signal P₁ is deasserted (andcontrol signal P₁ is asserted), top coils 102 a of inductors 100 a and100 b are in series to the power supply of battery 206 and ground. Theconfiguration shown in FIG. 10B is a transfer phase of augmentedtwo-stage boost converter 900 in which energy is transferred from topcoils 102 a to capacitor 204 and load 202.

Thus, the unique behavior of charging coils 102 a from the two stages inparallel and transferring stored energy from coils 102 a in series maybe an advantage of this architecture. The bottom coils 102 b ofinductors 100 a and 100 b may be controlled in a similar manner.

Because first stage 901 a and augmented second stage 901 b charge inparallel and transfer in series, the total boost voltage ratio is thesum of the contribution of each stage, as given by:

$\begin{matrix}{\frac{V_{out}}{V_{in}} = \frac{2}{1 - D}} & (4)\end{matrix}$

assuming no resistive losses. Equation 4 shows that the boost action ofeach stage 901 a, 901 b combines additively, in contrast to two-stageboost converter 400 in which the boost action of each stage 401 a, 401 bcombines multiplicatively. As a result, augmented two-stage boostconverter 900 may require a smaller duty cycle than single-stage boostconverter 200 in order to achieve the same boost ratio (though to alesser extent than two-stage converter 400) which may minimize currentripple.

When control signal P₁ of augmented two-stage boost converter 900transitions from asserted to deasserted (and control signal P₁transitions from deasserted to asserted), it is possible that currentsI_(1-STAGE1) and I_(1-STAGE2) in coils 102 a may not be exactly equal.This unequal current may occur because when control signal P₁ isasserted, the conduction path resistance for coils 102 a of inductors100 a and 100 b may be different (e.g., inductor 100 b may have an extraswitch in its conduction path when control signal P₁ is asserted thatcan add switch resistance). When two inductors with different currentsare connected in series, the current in one (or both) of the inductorsmust change rapidly to satisfy continuity. However, rapid changes ofcurrent in inductors may generate large, potentially damaging voltagesin the circuit. To solve this problem, switch 912 may couple capacitor905 between the common electrical node of coils 102 a (when controlsignal P₁ is deasserted) and ground, providing an alternative path toany such excess current. For bottom coils 102 b of inductors 100 a and100 b, switch 916 may be used for a similar purpose for which switch 912is used.

In contrast with capacitor 405 of two-stage boost converter 400,capacitor 905 may be much smaller with minimal impact to total circuitarea. In fact, in some instances such capacitor could have asufficiently small capacitance that capacitor 905 may be formed withinthe integrated circuit of augmented two-stage power converter 900. Anatural consequence of the architecture of augmented two-stage powerconverter 900 is that capacitor 905 may balance current between firststage 901 a and augmented second stage 901 b.

FIGS. 11A-11C depict a circuit simulation of currents for themulti-wound integrated inductors of the augmented two-stage boostconverter 900 shown in FIG. 9 over one pulse-width modulation cycle, inaccordance with embodiments of the present disclosure. FIG. 11A depictsexample control signals P₁ and P₂ and FIG. 11B depicts currents of coils102 a and 102 b of inductors 100 a and 100 b. During the states wheneither control signal P₁ is asserted and control signal P₂ is deassertedor control signal P₁ is deasserted and control signal P₂ is asserted, atleast of a pair of coils 102 is coupled in series to load 202 as shownin FIG. 10B. In these states, energy may be transferred from themagnetic fields of inductors 100A and 100B at the same time energy maybe simultaneously stored in the magnetic field. In effect, in suchstates, energy may be transferred from one coil 102 of an inductor 100to the other coil 102 of the inductor. Such transformer action may keepexcessive energy from building up in magnetic core 104, therebypotentially preventing early saturation.

Augmented two-stage power converter 900 may prevent current saturationbecause it may minimize the total magnetic field in magnetic core 104,thereby minimizing the amount of magnetic energy stored in magnetic core104. The total magnetic field in magnetic core 104 may be proportionalto magnetization current, I_(mag), which (for each inductor 100) may bedefined as:

I _(mag) =I ₁ −I ₂   (5)

When magnetization current I_(mag) is greater than or equal tomagnetization current saturation limit I_(diff) ^(sat), magnetic core104 may saturate.

FIG. 11C depicts magnetizing currents I_(mag1) and I_(mag2) forinductors 100 a and 100 b, respectively. When control signals P₁ and P₂are both asserted, currents in coils 102 are both increasing becauseboth are coupled in parallel between power supply and ground, aspreviously shown in FIG. 10A. However, during this state, themagnetizing currents I_(mag1) and I_(mag2) stay relatively flat becausethe flux generated by each coil 102 is changing at equal rates, and thusthe difference remains constant. This flatness of currents may, ineffect, create a “flat-top” to the magnetizing current waveforms asshown in FIG. 11C that prevents the magnetizing currents I_(mag1) andI_(mag2) from saturating.

The advantage of the two-stage, augmented boost architecture depicted inFIG. 9 is that it may reduce peak currents compared to the single-stagearchitecture depicted in FIG. 2 and the two-stage architecture depictedin FIG. 4 and may minimize energy stored in the core of inductors 100,thus minimizing a likelihood of core saturation. Because of theseadvantages, the current control methodology described below contemplatesuse of the two-stage, augmented boost architecture depicted in FIG. 9.However, the concepts, methods, and systems discussed below could beextended to the single-stage architecture depicted in FIG. 2, thetwo-stage architecture depicted in FIG. 4, or any other boostarchitectures, including without limitation those disclosed in U.S.patent application Ser. No. 16/692,072 filed Nov. 22, 2019, which isincorporated by reference herein in its entirety.

As discussed above, the output of augmented two-stage boost converter900 may be controlled by the duty cycles of control signals P₁ and P₂.These duty cycles may be time-varying signals that must be carefullychosen to prevent core saturation due to disturbances and regulate theamount of output current.

FIGS. 11A-11C discussed above depict idealized pulse-width modulationand current waveforms during steady-state operation. However, inreality, augmented two-stage boost converter 900 may experience somedeviations from these idealized waveforms. If the pulse-width modulatedcontrol signals P₁ and P₂ are generated by a digital process, then clockjitter, quantization effects, and mismatch in the digital components maycause the actual pulse-width modulated control signals P₁ and P₂ to varyfrom what is commanded. In addition, non-idealities in analog circuitryof augmented two-stage boost converter 900, such as switching time oftransistor switches, gate drive effects (e.g., switch non-overlap time),and thermal effects, may cause additional deviations in the response ofaugmented two-stage boost converter 900.

For example, FIG. 12A depicts an example model for modeling effects ofdisturbance in generation of pulse-width modulated control signals P₁and P₂ , and FIG. 12B depicts an ideal pulse-width modulated controlsignal (e.g., control signal P₁) and the ideal pulse-width modulatedcontrol signal affected by a disturbance, in accordance with embodimentsof the present disclosure. As shown in FIG. 12A, two separatepulse-width modulation (PWM) generators 1202 may use two independentduty cycles D₁ and D₂ to generate control signals P₁, P₂, P₁ , and P₂for augmented two-stage boost converter 900. Each PWM generator 1202 mayhave an independent disturbance source that causes a perturbation in itsgenerated PWM waveform, as shown in FIG. 12B. This disturbance may moveone or more edges of the PWM waveform, effectively changing its dutycycle.

Such disturbances may be problematic as they may lead to inductorsaturation. FIGS. 13A and 13B depict a simulation of an example stepdisturbance using the disturbance model of FIG. 12A. In the example ofFIGS. 13A and 13B, a 1% step disturbance is introduced in duty cycle D₁at a time t=0.2 μs, while duty cycle D₂ remains undisturbed as shown inFIG. 13A. FIG. 13B shows a simulation of currents through coils 102 aand 102 b of inductors 100 responsive to the step disturbance.

FIG. 14A depicts magnetizing currents I_(mage1) and I_(mag2) forinductors 100 a and 100 b, respectively, resulting from the response ofaugmented two-stage boost converter 900 to the step disturbance depictedin FIGS. 13A and 13B, along with saturation level I_(diff) ^(sat), inaccordance with embodiments of the present disclosure. FIG. 14Billustrates a saturation profile of currents within the multi-woundintegrated inductors of augmented two-stage boost converter 900 during acycle of operation prior to a disturbance in generation of a pulse-widthmodulated control signal, in accordance with embodiments of the presentdisclosure. Similarly, FIG. 14C illustrates a saturation profile ofcurrents within the multi-wound integrated inductors of augmentedtwo-stage boost converter 900 during a cycle of operation after adisturbance in generation of a pulse-width modulated control signal, inaccordance with embodiments of the present disclosure.

Before the disturbance of FIG. 13A is applied, as shown in FIG. 14B,currents of inductors 100 may remain within the saturation boundaryI^(sat). However, as shown in FIG. 14C, after a number of cycles afterthe disturbance, one or more currents of inductors 100 may exceed thesaturation boundary I^(sat).

Accordingly, as shown above, a 1% disturbance in control signals P₁, P₂,P₁ , and P₂ may be sufficient to cause a failure of augmented two-stageboost converter 900 due to inductor core saturation. Larger disturbancesin either or both of control signals P₁, P₂ may cause augmentedtwo-stage boost converter 900 to exceed its limits by an even greaterextent. Thus, augmented two-stage boost converter 900 may be verysensitive to disturbance generation of control signals P₁, P₂, P₁ , andP₂ . Such sensitivity may be largely due to the small inductance valuesof integrated inductors 100. To ensure proper operation, control signalsP₁, P₂, P₁ , and P₂ may be controlled to regulate currents and preventinductor saturation from disturbances.

In addition to the disturbance rejection problem discussed above,control signals P₁, P₂, P₁ , and P₂ may be controlled to deliver powerto load 202 of augmented two-stage boost converter 900 and maintain aregulated output boost voltage V_(out). In a boost converter for audioapplications, output boost voltage V_(out) may vary by three times ormore to meet the requirements of a class H or class G/H amplifier.Loading on the output of augmented two-stage boost converter 900 mayvary over a wide range as well. Audio content may have a high crestfactor and may vary rapidly from silence (0 Watts) to full scale (˜10Watts) in tens of microseconds. Additionally, where load 202 is aspeaker, impedance of load 202 may vary by two times or more acrossoperating frequencies and temperatures. Thus, it may be desirable thataugmented two-stage boost converter 900 be capable of responding torapidly changing output voltage commands and loading conditions.

A voltage control loop may be commonly used to meet the requirements oftime varying output voltages and loads. FIG. 15 illustrates selectedcomponents of an example voltage control loop 1500 that may be used inconnection with a boost converter 1506 (e.g., which may be implementedwith augmented two-stage boost converter 900 or any other suitable boostconverter), in accordance with embodiments of the present disclosure.

As shown in FIG. 15, a compensator 1502 may receive a reference voltageV_(ref) which may represent the desired regulated output boost voltageV_(out), and which may be time-varying. Compensator 1502 may comparereference voltage V_(ref) and sensed output boost voltage V_(out) andgenerate a control signal CTRL based on the comparison and a controlalgorithm. In many architectures, control signal CTRL may drive an innercurrent control loop implemented by a current controller 1504. Suchcurrent control loop may regulate an amount of average current deliveredto capacitor 204 and load 202, thereby regulating output boost voltageV_(out). However, for proper operation, current controller 1504 may needto simultaneously prevent inductor saturation.

Thus, it may be desirable that a boost converter (e.g., augmentedtwo-stage boost converter 900 or other boost converter 1506) using amulti-wound inductor 100 have a current controller 1504 that meets tworequirements: (a) current controller 1504 regulates currents ininductors 100 to prevent saturation from disturbances; and (b) currentcontroller 1504 regulates an output current driven to the load of boostconverter 1506 to meet the requirements of audio systems with timevarying boost voltages and loading.

FIG. 16A illustrates selected components of an example current controlscheme that may be used in connection with boost converter 1506, inaccordance with embodiments of the present disclosure. As shown in FIG.16A, current controller 1504 may include a measurement block 1602configured to receive a sensed inductor current I (e.g., a current I₁ orI₂ of an inductor 100) and perform a calculation on sensed inductorcurrent I to generate a signal that is indicative of inductor coilcurrents I₁ or I₂. Current controller 1504 may have two externalsignals: (a) saturation control signal SATCTRL which may regulatecurrent saturation; and (b) a current control signal ICTRL whichcontrols a target output current delivered to load 202. A control block1604 may receive an output of measurement block 1602, saturation controlsignal SATCTRL, and current control signal ICTRL to generate controlsignals for switches of boost converter 1506.

FIG. 16B illustrates selected components of an example control subsystem1600 that may be used in connection with boost converter 1506 and thecurrent control scheme depicted in FIG. 16A, in accordance withembodiments of the present disclosure. As shown in FIG. 16B, compensator1502 may be used to generate current control signal ICTRL based on acomparison of reference voltage V_(ref) and sensed output boost voltageV_(out), which may allow compensator 1502 to regulate an amount ofcurrent delivered from boost converter 1506 and thus regulate outputboost voltage V_(out). As shown in FIG. 16B, saturation control signalSATCTRL may be permanently set to zero to minimize energy stored in thecores of inductors 100 and prevent saturation due to disturbances.Control block 1604 may receive an output of measurement block 1602(indicative of measured current inductor coil currents I₁ or I₂),saturation control signal SATCTRL, and current control signal ICTRL togenerate control signals P₁, P₂, P₁ , and P₂ for switches of boostconverter 1506.

The discussion below outlines several example embodiments, grouped intofour sections related to either control or measurement of circuitparameters, for implementing control subsystem 1600.

1. Generating PWM Control Signals and Calculating Cycle Averages

The measurement process of measurement block 1602 may comprisecalculating the cycle averages of inductor coil currents I₁ and I₂ ofcoils 102 a and 102 b of inductors 100 in a manner that is coordinatedor linked to the PWM generation process. FIG. 17A illustrates selectedcomponents, including a cycle average calculator 1704, for performingcalculation of cycle averages of inductor coil currents I₁ and I₂, inaccordance with embodiments of the present disclosure.

To further illustrate, such functionality, FIGS. 18A and 18B depict asimulation of an example 1% step disturbance introduced in duty cycle D₁at a time t=0.2 μs, while duty cycle D₂ remains undisturbed as shown inFIG. 18A. FIG. 18B shows a simulation of currents through coils 102 aand 102 b of a dual-wound inductor 100 responsive to the stepdisturbance. FIG. 18C depicts a plot of the average winding currents I₁and I₂ of coils 102 a and 102 b. Calculating an average of currents I₁and I₂ may remove the effect of the current ripple within a PWM cycleand may be a much clearer indicator of the current dynamic behavior. Theuse of average currents may therefore simplify current control bycurrent controller 1504.

Notably, FIGS. 18B and 18C depict only currents I₁ and I₂ of coils 102 aand 102 b of first-stage inductor 100 a of augmented two-stage powerconverter 900 and omit currents for second-stage inductor 100 b.However, because of the augmented two-stage architecture, currents I₁and I₂ for inductor 100 a may be similar to that of 100 b, and currentcontroller 1504 may take advantage of this similarity by only measuringcurrents of one of inductors 100 and controlling all currents based onsuch measurement.

Cycle average calculator 1704 may calculate average cycle values forcurrents I₁ and I₂ in any suitable manner, including directly bysampling multiple points and summing, implementing a circuit thatautomatically integrates currents I₁ and I₂ over a cycle, or using atriangle carrier signal of a triangle modulator 1702 to generatepulse-width modulated control signals P₁, P₂, P₁ , and P₂ , as is knownin the art, and use such triangle carrier signal to trigger a midpointsampler 1706 configured to sample values of currents I₁ and I₂ at themidpoint of the “ON” time of a pulse-width modulated control signals P₁,P₂, as depicted in FIG. 17B. Midpoint sampler 1706 may provide anapproximation of cycle average of currents I₁ and I₂, as described ingreater detail below.

Midpoint sampler 1706 may sample both currents I₁ and I₂ simultaneouslyor perform an alternate sampling of currents I₁ and I₂ as shown in FIG.17C. FIG. 17C shows a portion of augmented two-stage boost converter 900in which switches 210 and 214 are controlled by control signals P₁ andP₂, respectively. In the embodiments represented by FIG. 17C, midpointsampler 1706 may be implemented by sense resistor 1708 andanalog-to-digital converter (ADC) 1710. Sense resistor 1708 may becoupled between a ground voltage and a common node of switches 210 and214. ADC 1710 may sample a voltage across sense resistor 1708 which maybe indicative of a current flowing to ground. Analog-to-digitalconverter 1710 may be triggered by the trigger signal from trianglemodulator 1702 to sample a midpoint current value. Using a single ADC1710 and sense resistor 1708 may be advantageous because they make up asmaller circuit and the smaller circuit may avoid possible mismatch iftwo ADCs and sense resistors were to be used.

FIGS. 19A-19C depict various waveforms for a three-cycle simulation ofcoordination of midpoint sampler 1706 with triangle modulator 1702 andthe sampling process of FIG. 17C, in accordance with embodiments of thepresent disclosure. FIGS. 19A and 19B depict generation of pulse-widthmodulated control signals P₁ and P₂ from a triangle carrier wave. FIG.19A depicts a triangle carrier wave CARRIER with a minimum of −1 and amaximum of +1 and a period equal to a pulse-width modulation period ofpulse-width modulated control signals P₁ and P₂. Reference signals ref1and ref2 may be related to the desired duty cycles D₁ and D₂ as follows:

ref1=2D ₁−1   (5)

ref2=1−2D ₂   (6)

FIG. 19B depicts pulse-width modulated control signals P₁ and P₂ whichmay be derived from reference signals ref1 and ref2 and triangle carrierwave CARRIER as follows:

$\begin{matrix}{P_{1} = \left\{ \begin{matrix}{1,{{{ref}\; 1} > {CARRIER}}} \\{0,{otherwise}}\end{matrix} \right.} & (7) \\{P_{2} = \left\{ \begin{matrix}{1,{{{ref}\; 2} < {CARRIER}}} \\{0,{otherwise}}\end{matrix} \right.} & (8)\end{matrix}$

Control signals P₁ and P₂ may be the logical complements of controlsignals P₁ and P₂, respectively.

The alternate sampling of currents I₁ and I₂ as shown in FIG. 17C may beachieved when triangle carrier wave CARRIER equals −1 and +1,respectively. The times of alternate sampling of currents I₁ and I₂ aredepicted in FIGS. 19A-19C by points labeled I₁samp and I₂samp,respectively. By sampling at the maximum and minimum of triangle carrierwave CARRIER, sampling is configured to occur at the midpoint of the“ON” time of control signals P₁ and P₂. Such midpoint sampling may beadvantageous as it is maximally far away in tome from the edges ofcontrol signals P₁ and P₂ where switching transients may distortmeasurement. In addition, midpoint sampling may ensure that current I₁is sensed when control signal P₁ is asserted and control signal P₂ isdeasserted and that current I₂ is sensed when control signal P₂ isasserted and control signal P₁ is deasserted, regardless of the desiredduty cycles D₁ and D₂.

Sampling at the midpoint of the PWM waveform has the additional benefitthat it may approximate the cycle-average of the inductor currents I₁and I₂. FIG. 19C depicts simulated winding currents I₁ and I₂ assuminglossless inductors and switches, along with the sampled current values.Due to the symmetry of the switching states and the accompanyinganti-symmetry of the currents about the midpoint of the control signal“ON” time, the sampled values may represent cycle averages of inductorcurrents I₁ and I₂.

During real operation, resistive losses along with imbalance between twostages 901 a and 901 b of augmented two-stage boost converter 900 maydistort current waveforms such that they are no longer piecewise-linearwaveforms that are antisymmetric about the midpoint of the “ON” times ofcontrol signals P₁ and P₂. This distortion may cause an offset betweenthe actual cycle-averages of inductor currents I₁ and I₂ and the valuesobtained from mid-point sampling. FIGS. 20A and 20B show an example ofthis offset from a simulation with resistive losses and imbalancebetween stages 901 a and 901 b.

Despite these offset errors, the midpoint-sampled values of inductorcurrents I₁ and I₂ may still be used in the current control loopdescribed above. First, the errors are generally small relative to thevalue being measured. Second, the outer voltage regulation loop may usean integrator to zero out any error in the commanded output current.Third, as the winding current difference approaches zero, the estimatedwinding current difference from midpoint sampling also approaches zero.Therefore, if estimated coil current difference |I₁−I₂| is controlled tozero, the actual winding current difference |1 ₁−I₂| may also approachzero which may prevent saturation.

2. Applying a Transformation to Current Measurement and Duty Cycle

In some embodiments, the control subsystem of boost converter 1506 mayimplement a coordinate transformation to decouple signals in order tosimplify control of boost converter 1506. FIGS. 21A and 21B illustrateselected components, including a forward transform block 2102 and aninverse transform block 2104, for coordinate transformation, inaccordance with embodiments of the present disclosure. As shown in FIGS.21A and 21B, forward transform block 2102 may be applied to measurementsfor inductor currents I₁ and I₂ and control block 1604 may operate ontransformed measurements I_(m) and I_(b) for inductor currents I₁ and I₂to generate duty cycle control signals D_(m) and D_(b) which may in turnbe inverse transformed by inverse transform block 2104 to generate dutycycle control signals D₁ and D₂ used to drive PWM generator 1202.

Forward transform block 2102 may apply the following transform togenerate transformed current measurements I_(m) and I_(b):

$\begin{matrix}{\begin{bmatrix}I_{m} \\I_{b}\end{bmatrix} = {\begin{bmatrix}1 & {- 1} \\1 & 1\end{bmatrix} \cdot \begin{bmatrix}I_{1} \\I_{2}\end{bmatrix}}} & (9)\end{matrix}$

wherein I_(m) may be referred to as a magnetizing current and I_(b) maybe referred to as a battery current. Magnetizing current may be equal tothe difference between inductor currents I₁ and I₂ and may therefore beproportional to the net magnetic field in the core of multi-woundinductor 100. The term “battery current” as used herein is not in anyway limited to a current sourced from a battery but may be sourced froma battery or any suitable power supply or may be a mathematicalequivalent/transformative value representative of a battery current orpower supply current. For example, battery current I_(b) may be the sumof coil currents I₁ and I₂ and for a single stage converter, batterycurrent I_(b) may be exactly equal to an actual current flowing from thebattery or power supply. For a two-stage converter, battery currentI_(b) may no longer be equal to an actual battery or power sourcecurrent, but may be so termed because it is the same mathematicaltransformation. Because coil currents I₁ and I₂ may be the cycle averagevalues of the winding currents, magnetizing current I_(m) and batterycurrent I_(b) may also represent the cycle averages of the magnetizingand battery currents.

The inverse transform of transform block 2102 may be applied by inversetransform block 2104 to generate duty cycle control signals:

$\begin{matrix}{\begin{bmatrix}D_{1} \\D_{2}\end{bmatrix} = {{\frac{1}{2}\begin{bmatrix}1 & 1 \\{- 1} & 1\end{bmatrix}} \cdot \begin{bmatrix}D_{m} \\D_{b}\end{bmatrix}}} & (10)\end{matrix}$

FIG. 21B depicts how both transformations are implemented. In this case,the saturation control signal SATCTRL and the current control signalICTRL received by control block 1604 may be replaced by a magnetizingcurrent reference I_(m) ^(ref) and a battery current reference I_(b)^(ref), respectively. Because of this transformation, control block 1604may operate in magnetizing/battery coordinate space rather than a coil101 a/coil 102 b coordinate space.

FIG. 21C illustrates a saturation profile of currents within amulti-wound integrated inductor 100 depicting transformed coordinateaxes, in accordance with embodiments of the present disclosure. The plotof FIG. 21C depicts a saturation region 110 similar to that of FIG. 1Con a coordinate axis in which current I₁ is on the horizontal axis andcurrent I₂ is on the vertical axis. The transformation of equation (9)may be interpreted as a rotation transformation that rotates the (I₁,I₂) axes by 45° to a new set of axes (I_(m), I_(b)) as shown in FIG.21C. This rotation transformation may be useful because for most of theunsaturated region 112, the magnetizing current coordinate may be adirect measurement of how far away boost converter 1506 is fromexceeding the magnetizing saturation limit I_(diff) ^(sat). On the otherhand, the orthogonal battery current I_(b) coordinate is free from thisconstraint and may be representative of a current flowing through theboost converter 1506 to its output. At large current values, batterycurrent I_(b) may have a maximum current constraint, but theapproximation in the transformed coordinate space may be useful forcontrol over most of the operating space. This example embodimentdepicts how the transformation may decouple the saturation protectionfrom output current requirements, allowing each quantity to becontrolled independently.

The transformations of equations (9) and (10) may also have the addedbenefit of decoupling the system modes of the converter. Augmentedtwo-stage boost converter 900 may be modeled using the examplestate-space averaging technique described in Erickson, Robert W. andDragan Maksimociv, “Fundamentals of Power Electronics: Second Edition,”Springer Science+Business Media, 2001, which is incorporated herein byreference. Such model may also be linearized about a nominal operatingpoint using a small-signal approximation. Assuming the current dynamicsare much faster than the output voltage dynamics, the system includingboost converter 1506 and its control subsystem may be modeled as asecond-order ordinary differential equation as follows:

$\begin{matrix}{{\begin{bmatrix}L & M \\{- M} & L\end{bmatrix} \cdot {\frac{d}{dt}\begin{bmatrix}i_{1} \\i_{2}\end{bmatrix}}} = {{\begin{bmatrix}{- \hat{R}} & 0 \\0 & {- \hat{R}}\end{bmatrix} \cdot \begin{bmatrix}i_{1} \\i_{2}\end{bmatrix}} + {\begin{bmatrix}\hat{V} & 0 \\0 & \hat{V}\end{bmatrix} \cdot \begin{bmatrix}d_{1} \\d_{2}\end{bmatrix}}}} & (11)\end{matrix}$

where lower case variables i₁, i₂, d₁, and d₂ represent the small signaldeviations of currents I₁ and I₂ and duty cycles D₁ and D₂ fromsteady-state, L is the self-inductance of inductor 100, M is the mutualinductance of inductor 100, {circumflex over (R)} is a resistance whichis a function of a switch resistance, inductor resistance, andsteady-state duty cycle, and {circumflex over (V)} is a voltage that isa function of the power supply voltage, output boost voltage V_(out),switch resistance, inductor resistance, and steady-state current. If thetransformations of equations of (9) and (10) are applied to equation(11), the result may be:

$\begin{matrix}{{{L\begin{bmatrix}{1 + k} & 0 \\0 & {1 - k}\end{bmatrix}} \cdot {\frac{d}{dt}\begin{bmatrix}i_{m} \\i_{b}\end{bmatrix}}} = {{\begin{bmatrix}{- \hat{R}} & 0 \\0 & {- \hat{R}}\end{bmatrix} \cdot \begin{bmatrix}i_{m} \\i_{b}\end{bmatrix}} + {\begin{bmatrix}\hat{V} & 0 \\0 & \hat{V}\end{bmatrix} \cdot \begin{bmatrix}d_{m} \\d_{b}\end{bmatrix}}}} & (12)\end{matrix}$

where lower case variables i_(m), i_(b), d_(m), and d_(b) represent thesmall signal deviations of currents I_(m) and I_(b) and duty cyclesD_(m) and D_(b) from steady-state, and k is a coupling coefficientdefined by M/L. In this case, the matrices are diagonalized, such thatthe magnetizing and battery modes are orthogonal. Thus, the system ofboost converter 1506 and its control subsystem may be decoupled intotwo, independent first-order modes.

FIGS. 22A-22D depict the results when the foregoing transformation isapplied to the previous example of a 1% step disturbance in duty cycleD₁. FIG. 22A depicts duty cycle D₁ with a disturbance occurring at 0.2μsec, and FIG. 22B depicts the simulated cycle-average of coil currentsI₁ and I₂ for inductor 100 a. Even though the disturbance occurs only onduty cycle D₁, both currents I₁ and I₂ may be affected due to thecoupled dynamics of equation (11). Transformation equation (9) may beapplied to these currents I₁ and I₂ to yield magnetizing current I_(m)in FIG. 22C and battery current I_(b) in FIG. 22D. Currents I₁ and I₂may exhibit a second-order response whereas magnetizing current I_(m)and battery current I_(b) may exhibit a simple first-order response.These first-order, decoupled dynamics of the magnetizing current I_(m)and battery current I_(b) may be easier to control within a controlsubsystem.

3. Using an Observer for Current Measurement

The cycle-average measurements of currents I₁ and I₂ may be noisy.Additionally, if the sampling method from FIG. 17C is implemented,information from one of the coils 101 may be missing at each samplingperiod. These shortcomings may degrade the performance of the controlsubsystem. Accordingly, the control subsystem may implement an observer2300, as shown in FIG. 23A. In operation, observer 2300 may receivemeasured current data I_(meas) and generate an improved estimate ofcurrents I₁ and I₂. Observer 2300 may use a model of the controlsubsystem to filter out noise and fill in any missing information. Thesystem model may require knowledge of or information related to thedesired duty cycles D₁ and D₂, as well as an indicator of which currentvalue I₁ and I₂ is currently being measured. The latter indicator signalis labeled I_(meas) mode in FIG. 23A. Observer 2300 may be implementedusing one of several methodologies including, without limitation: aLuenberger filter, a Kalman filter, and a slide mode observer. FIG. 23Bshows an implementation based on a Kalman filter. The Kalman filterarchitecture of FIG. 23B may be used if measured current data I_(meas)comes from the current sampling scheme shown in FIG. 17C. The Kalmanfilter implementation of FIG. 23B may operate by using a model 2302 andthe known model inputs of desired duty cycles D₁ and D₂ to form anestimate of currents I₁ and I₂. These estimates may be compared by asubtractor 2304 with measured current data I_(meas) to form error signale. Error signal e may be multiplied by Kalman gains, K, and then used toadjust the model estimates such that the error is minimized. The signalI_(meas) mode may indicate whether measured current data I_(meas) ismeasuring current I₁ or current I₂.

The system model may be derived by discretizing the continuous-timemodel equation (11) using any standard method (e.g., forward Euler,bilinear transform, Zero Order Hold (ZOH), etc.) and re-writing it instandard form:

$\begin{matrix}{\begin{bmatrix}i_{1} \\i_{2}\end{bmatrix}_{i + 1} = {{A \cdot \begin{bmatrix}i_{1} \\i_{2}\end{bmatrix}_{i}} + {B \cdot \begin{bmatrix}d_{1} \\d_{2}\end{bmatrix}_{i}}}} & (13)\end{matrix}$

Where A and B are 2×2 matrices and I is the sample time index. Thismodel may be extended to include the effects of noise and disturbanceson generation of control signals P₁ and P₂:

$\begin{matrix}{x_{i + 1} = {{\underset{\underset{F}{︸}}{\begin{bmatrix}A & B \\0 & I\end{bmatrix}}x_{i}} + {\underset{\underset{G}{︸}}{\begin{bmatrix}B \\0\end{bmatrix}}u_{i}} + w_{i}}} & (14) \\{y_{i} = {{H_{i}x_{i}} + r_{i}}} & (15)\end{matrix}$

where x_(i)=[i₁ i₂ dist1 dist2]_(i) ^(T) is the state vector and dist1and dist2 are disturbance estimates; u_(i)=[d₁ d₂ 0 0]_(i) ^(T) is themodel input vector; F is a 4×4 block matrix comprising the 2×2 matrix Afrom equation (13), the 2×2 matrix B from equation (13), the 2×2 zeromatrix, and the 2×2 identity matrix; G is a 4×2 block matrix comprising2×2 matrix B and the 2×2 zero matrix, w is a 4×1 vector of the processnoise, y is the output I₁ or I₂; and r is the scalar measurement noise.Matrix H_(i) may change with time depending on the value of signalI_(meas) mode: if current I₁ is being measured, H_(i)=[1 0 0 0] and ifcurrent I₂ is being measured, H_(i)=[0 1 0 0]. The dependency of model2302 on signal I_(meas) mode is represented in FIG. 23B with a dashedline.

The Kalman filter implemented by model 2302 may be implemented in anysuitable number of ways using equations (14) and (15) (e.g., inaccordance with Simon, Dan, “Optimal State Estimation,” Wiley 2006,which is incorporated by reference herein in its entirety). In onepossible implementation, the Kalman recursion may be given by:

z _(i) =Fx _(i−1) ^(e) +Gu _(i−1)   (16)

x _(i−1) ^(e) =z _(i) +K _(i)·(y _(i) −H _(i) z _(i))   (17)

where x_(i−1) ^(e)=[i₁ ^(e) i₂ ³ dist1 ^(e) dist2 ^(e)]_(i) ^(T) is theestimated state vector that contains estimated coil currents anddisturbances and z_(i) is an internal state vector (e.g., an a prioristate estimate).

The Kalman gain, K_(i), may be derived using any of the standardtechniques (e.g., techniques disclosed in the Simon reference citedabove). However, the gain computation must account for the H_(i) matrixthat changes with time based on measured current data I_(meas). In oneimplementation, the Kalman gains may be calculated ahead of time overseveral samples. At steady state, the Kalman gains will alternatebetween two sets of values depending on signal I_(meas) mode. Thesesteady-state values may be stored and applied depending on the state ofsignal I_(meas) mode, as indicated by the dashed lines in FIG. 23B.

FIGS. 24A-24D depict simulation results of augmented two-stage powerconverter 900 in response to the 1% step disturbance earlier described.The Kalman filter of FIG. 23B may be used to estimate the currents I₁and I₂ as well as the disturbances on control signals P₁ and P₂. Thesimulation shown may use the current sampling technique disclosed inFIG. 17C. FIGS. 24A and 24B depict actual and estimated disturbances oncontrol signals P₁ and P₂ while FIGS. 24C and 24D depict actual andestimated currents I₁ and I₂. The currents plotted may be the deviationsfrom steady state. The Kalman filter of FIG. 23B may rapidly converge onan estimate of both the disturbances of control signals P₁ and P₂ andcurrents I₁ and I₂ even though the measured value of measured currentdata I_(meas) only includes one of currents I₁ and I₂ on each sample.

FIG. 23C depicts an alternative embodiment for observer 2300. In theembodiment of FIG. 23C, two independent observers each with a differencemodel 2302 a and 2302 b and gain K₁ and K₂ may be used. First observer2300 a may form an estimate of current I₁ when current I₂ is beingmeasured. Its input may be the instant measurement of current I₁ and theprevious measurement of current I₂. Likewise, second observer 2300 b mayform an estimate of current I₂ when current I₁ is being measured. Itsinput may be the instant measurement of current I₂ and the previousmeasurement of current I₁. Unlike the example of FIG. 23B, models 2302a, 2302 b and gains K₁, K₂ may be fixed. Estimates for current I₁, I₂may be taken from either the output of first model 2302 a or secondmodel 2302 b depending on the state of signal I _(meas) mode which maycontrol a multiplexed output 2306.

4. Applying Independent Control on Decoupled Current Signals

As shown in FIG. 25A, in some embodiments, currents I₁ and I₂ may becontrolled using two independent control loops with control blocks 1604a and 1604 b that separately control the magnetizing and battery modes.Control blocks 1604 a and 1604 b may be implemented using any standardcontrol algorithm including without limitation proportional,proportional-integral, proportional-integral-derivative, or state-space.FIG. 25B depicts an example of a state-space control 2500 that may beused for either or both of control blocks 1604 a and 1604 b. State-spacecontrol 2500 may implement a servo control architecture that includes anadded integrator 2502 to remove steady-state error (e.g., as disclosedin Ogata, Katsuhiko, Discrete-Time Control Systems, Prentice Hall,1995). Such control may be based on a decoupled state-space model of thesystem. The decoupled model may be derived by applying themagnetizing/battery transform equations (9) and (10) to the discretizedmodel of equation (13):

$\begin{matrix}{\begin{bmatrix}i_{m} \\i_{b}\end{bmatrix}_{i + 1} = {{\begin{bmatrix}a_{m} & 0 \\0 & a_{b}\end{bmatrix} \cdot \begin{bmatrix}i_{m} \\i_{b}\end{bmatrix}_{i}} + {\begin{bmatrix}b_{m} & 0 \\0 & b_{b}\end{bmatrix} \cdot \begin{bmatrix}d_{m} \\d_{b}\end{bmatrix}_{i}}}} & (18)\end{matrix}$

Where a_(m), a_(b), b_(m), and b_(b) are scalar coefficients. Applyingthe magnetizing/battery transforms may diagonalize the matrices anddecouple the magnetizing/battery modes as discussed earlier. As aresult, independent state-space models may be written for each mode. Ina real implementation, there may be a non-zero computational delay timefor all the blocks of the algorithm. However, in this case, it isassumed there is a one-sample delay between receiving measured currentdata I_(meas) and calculating the next PWM command This one-sample delaymay also be included in the model for each mode as follows:

$\begin{matrix}{\begin{bmatrix}{up} \\i_{m}\end{bmatrix}_{i + 1} = {{\begin{bmatrix}0 & 0 \\1 & a_{m}\end{bmatrix} \cdot \begin{bmatrix}{up} \\i_{m}\end{bmatrix}_{i}} + {\begin{bmatrix}b_{m} \\0\end{bmatrix}d_{m_{i}}}}} & (19) \\{\begin{bmatrix}{up} \\i_{b}\end{bmatrix}_{i + 1} = {{\begin{bmatrix}0 & 0 \\1 & a_{b}\end{bmatrix} \cdot \begin{bmatrix}{up} \\i_{b}\end{bmatrix}_{i}} + {\begin{bmatrix}b_{b} \\0\end{bmatrix}d_{b_{i}}}}} & (20)\end{matrix}$

where up is an internal state that represents the input during theprevious sample. The models given by equations (18) and (19) may each beused to construct a servo control using a method similar to the Ogatareference cited above. The result may be the architecture in FIG. 25B.The block 2504 labeled “b” may calculate a gain using either the b_(m)or b_(b) coefficient from equation (19) or (20) depending on whether thecontrol is for the magnetizing or battery mode. The gains K₁, K₂, and K₃may be calculated using any standard state-space method (e.g., poleplacement, quadratic optimal control, etc.).

The systems and methods described herein may only function successfullywhen coils 102 a and 102 b of a dual-wound inductor 100 having acoupling coefficient k within a particular range. Specifically, a valueof k satisfying 0.70≤k≤0.95 may be necessary for successful operation.First, for values of k outside such range, an open loop circuit willhave winding currents that can either saturate the inductor or haveexcessively large ripple. Second, for values of k outside this range, aclosed loop system regulating inductor current is difficult to control,impractical to implement, or unstable. Further, it is impractical orexpensive to manufacture embedded inductors with coupling coefficientsoutside this range.

Open-Loop Circuit Behavior

Coupling coefficient k must be above a minimum value for the circuitsdescribed above to function properly. In the Background and FIGS. 1A-1Cit was shown that coupling was necessary for the anti-wound coils 102 aand 102 b to cancel the magnetic field in the core and preventsaturation. Specifically, such coupling keeps the difference in windingcurrents below saturation level I_(diff) ^(sat) as shown in FIG. 3B andequation (1). If coupling coefficient k is too low, field cancellationwill not occur and condition (1) may be violated.

If the coupling coefficient is too high, a circuit comprisingmulti-wound inductor 100 will also not function properly. If couplingcoefficient k approaches 1, multi-wound inductor 100 will have too muchfield cancellation when the winding currents of the anti-wound inductorare both rising or falling. For example, as shown in FIGS. 11A and 11B,around times 1×10⁻⁸ seconds and 3×10⁻⁸ seconds, both gate controlsignals are positive, and all the winding currents are simultaneouslyrising. As coupling coefficient k approaches 1, the slope of the windingcurrents in these regions will approach infinity due to the fieldcancellation. This cancellation may cause high current ripple that candrive inductor 100 into the saturation region, shown in FIG. 1C, andpotentially damage the part. High current ripple can also causeexcessive heating in external components, generate radiation thatviolates electromagnetic compatibility (EMC) constraints, andpotentially damage a battery supply. Simulations have shown that for acoupling factor k between 0.70 and 0.95, the current ripple issufficiently small and the inductor winding currents stay in theunsaturated region.

Closed-Loop Circuit Behavior

Maintaining coupling coefficient k below a maximum limit is alsoimportant for closed-loop control that regulates electrical currentthrough inductor 100. A linearized model of voltage control loop 1500was given in Equation (12) using the magnetizing and battery currents asstate variables. Because the state variables are decoupled, theequations can be rearranged as two, independent, first-order ordinarydifferential equations:

$\begin{matrix}{\frac{{di}_{m}}{dt} = {{{- \frac{\hat{R}}{L \cdot \left( {1 + k} \right)}}i_{m}} + {\frac{\hat{V}}{L \cdot \left( {1 + k} \right)} \cdot d_{m}}}} & (21) \\{\frac{{di}_{b}}{dt} = {{{- \frac{\hat{R}}{L \cdot \left( {1 - k} \right)}}i_{b}} + {\frac{\hat{V}}{L \cdot \left( {1 - k} \right)} \cdot d_{b}}}} & (22)\end{matrix}$

In this form, it is clear that if coupling coefficient k approaches 1,the denominator of both coefficients in Equation (22), representingbattery current i_(b), may become very large. Such large coefficients inbattery current Equation (22) may be highly problematic for control. Ifk=1, then the coefficients in battery current Equation (22) are infiniteand the closed-loop system of voltage control loop 1500 becomes unstableand uncontrollable. However, even if coupling coefficient k approaches1, the closed-loop system of voltage control loop 1500 is difficult orimpractical to control. As the coefficient

$\frac{\hat{V}}{L \cdot \left( {1 - k} \right)}$

increases, the gain between the input signal d_(b) and the rate of rangeof current i_(b) may become large. In a practical implementation of acircuit having voltage control loop 1500, this extremely large gain maylead to undesirable amplification of noise from the input (e.g.,discretization noise) as well as lead to impractical demands forhigh-resolution control that can be costly. It can also cause largesignal swings on the currents that may saturate feedback sensors.

In addition, as the coefficient

$\frac{\hat{R}}{L \cdot \left( {1 - k} \right)}$

increase, the eigenvalue of Equation (22) may become large and currenti_(b) may have a very fast response. Systems with fast dynamics oftenrequire fast control circuitry which can be both costly to implement anduse large amounts of power, which decreases overall system efficiency.The fast response time of such a plant also means that it may be moresensitive to delays in the control loop, which also makes the controlsystem design more challenging.

Simulations performed by the Applicant have shown that constrainingk≤0.95 maintains the coefficients of Equation (22) small enough that acontrol circuit for voltage control loop 1500 may be practicallyimplemented.

Integrated, Dual-Inductor Manufacturability

Selecting a coupling coefficient k may also be important to keepmanufacturing costs of the integrated, dual-inductor 100 to a minimum aswell as enable successful integration into a target application.Coupling between coils 102 a and 102 b may be achieved by interleavingcoils 102 a and 102 b as shown in FIGS. 1A-1B. Coupling coefficient kmay be controlled by adjusting the number of turns of each coil 102 aand 102 b that are interleaved and the spacing between coils. Ifcoupling coefficient k is low, it means the two coils 102 and 102b mayneed to be further apart, which may increase the size and cost of adevice.

A coupling coefficient k that is too large is also impractical tomanufacture. Even with all the turns of both coils 102 a and 102 b fullyinterleaved, coupling coefficient k will reach a maximum value lessthan 1. Finite element simulations have shown such maximum value to bearound 0.95. This maximum value is because only some of the magneticflux generated by a coil 102 a/102 b flows through the core and linksthe adjacent coil 102 a/102 b. This linking is especially true for theend turns where there is less core material. Further increase incoupling coefficient k may be possible by adding more core materialaround the windings, but such approach may become very costly andimpractical to manufacture. In addition, the wiring to the coilstructure, both internal to an integrated circuit chip and external tothe chip, may have its own self-inductance. This factor alone mayprevent a coupling coefficient k greater than 0.99.

As used herein, when two or more elements are referred to as “coupled”to one another, such term indicates that such two or more elements arein electronic communication or mechanical communication, as applicable,whether connected indirectly or directly, with or without interveningelements.

This disclosure encompasses all changes, substitutions, variations,alterations, and modifications to the example embodiments herein that aperson having ordinary skill in the art would comprehend. Similarly,where appropriate, the appended claims encompass all changes,substitutions, variations, alterations, and modifications to the exampleembodiments herein that a person having ordinary skill in the art wouldcomprehend. Moreover, reference in the appended claims to an apparatusor system or a component of an apparatus or system being adapted to,arranged to, capable of, configured to, enabled to, operable to, oroperative to perform a particular function encompasses that apparatus,system, or component, whether or not it or that particular function isactivated, turned on, or unlocked, as long as that apparatus, system, orcomponent is so adapted, arranged, capable, configured, enabled,operable, or operative. Accordingly, modifications, additions, oromissions may be made to the systems, apparatuses, and methods describedherein without departing from the scope of the disclosure. For example,the components of the systems and apparatuses may be integrated orseparated. Moreover, the operations of the systems and apparatusesdisclosed herein may be performed by more, fewer, or other componentsand the methods described may include more, fewer, or other steps.Additionally, steps may be performed in any suitable order. As used inthis document, “each” refers to each member of a set or each member of asubset of a set.

Although exemplary embodiments are illustrated in the figures anddescribed below, the principles of the present disclosure may beimplemented using any number of techniques, whether currently known ornot. The present disclosure should in no way be limited to the exemplaryimplementations and techniques illustrated in the drawings and describedabove.

Unless otherwise specifically noted, articles depicted in the drawingsare not necessarily drawn to scale.

All examples and conditional language recited herein are intended forpedagogical objects to aid the reader in understanding the disclosureand the concepts contributed by the inventor to furthering the art, andare construed as being without limitation to such specifically recitedexamples and conditions. Although embodiments of the present disclosurehave been described in detail, it should be understood that variouschanges, substitutions, and alterations could be made hereto withoutdeparting from the spirit and scope of the disclosure.

Although specific advantages have been enumerated above, variousembodiments may include some, none, or all of the enumerated advantages.Additionally, other technical advantages may become readily apparent toone of ordinary skill in the art after review of the foregoing figuresand description.

To aid the Patent Office and any readers of any patent issued on thisapplication in interpreting the claims appended hereto, applicants wishto note that they do not intend any of the appended claims or claimelements to invoke 35 U.S.C. § 112(f) unless the words “means for” or“step for” are explicitly used in the particular claim.

What is claimed is:
 1. A system, comprising: a power convertercomprising at least one stage having a dual anti-wound inductor having afirst winding and a second winding constructed such that its windingsgenerate opposing magnetic fields in its magnetic core and constructedsuch that a coupling coefficient between the first winding and thesecond winding is less than approximately 0.95; and a current controlsubsystem for controlling an electrical current through the dualanti-wound inductor, the current control subsystem configured to:minimize a magnitude of a magnetizing electrical current of the dualanti-wound inductor to prevent core saturation of the dual anti-woundinductor; and regulate an amount of output electrical current deliveredby the power converter to the load in accordance with a reference inputsignal.
 2. The system of claim 1, wherein the power converter is a boostconverter.
 3. The system of claim 1, wherein the reference input signalis a reference voltage indicating a desired voltage level to be drivento the load.
 4. The system of claim 1, wherein the current controlsubsystem is further configured to control the magnetizing electricalcurrent and the output electrical current independently from oneanother.
 5. The system of claim 1, wherein the current control subsystemis further configured to, in order to minimize the magnitude of themagnetizing electrical current to prevent core saturation of the dualanti-wound inductor and regulate the amount of output electricalcurrent, transform electrical current parameters from a first coordinatespace defined by a first electrical current through a first winding ofthe dual anti-wound inductor and a second electrical current through asecond winding of the dual anti-wound inductor to a second coordinatespace defined by the magnetizing electrical current and the outputelectrical current.
 6. The system of claim 1, wherein the currentcontrol subsystem comprises an observer configured to improve accuracyof a measurement of a first electrical current through a first windingof the dual anti-wound inductor and a second electrical current througha second winding of the dual anti-wound inductor by estimating the firstelectrical current when the first electrical current is out of phase formeasurement and by estimating the second electrical current when thesecond electrical current is out of phase for measurement.
 7. The systemof claim 1, wherein the current control subsystem comprises a currentcontrol loop for control of the power converter, and is furtherconfigured to measure, as feedback parameters to the current controlloop: a first average current of a first winding of the dual anti-woundinductor over the duration of a pulse-width modulation period of thefirst winding; and a second average current of a second winding of thedual anti-wound inductor over the duration of a pulse-width modulationperiod of the second winding.
 8. The system of claim 1, wherein thefirst winding and the second winding are constructed such that thecoupling coefficient is greater than approximately 0.70.
 9. A method forcontrolling an electrical current through a dual anti-wound inductorintegral to a power converter, the dual anti-wound inductor having afirst winding and a second winding constructed such that its windingsgenerate opposing magnetic fields in its magnetic core and constructedsuch that a coupling coefficient between the first winding and thesecond winding is less than approximately 0.95, the method comprising:minimizing a magnitude of a magnetizing electrical current of the dualanti-wound inductor to prevent core saturation of the dual anti-woundinductor; and regulating an amount of output electrical currentdelivered by the power converter to the load in accordance with areference input signal.
 10. The method of claim 9, wherein the powerconverter is a boost converter.
 11. The method of claim 9, wherein thereference input signal is a reference voltage indicating a desiredvoltage level to be driven to the load.
 12. The method of claim 9,further comprising controlling the magnetizing electrical current andthe output electrical current independently from one another.
 13. Themethod of claim 9, further comprising, in order to minimize themagnitude of the magnetizing electrical current to prevent coresaturation of the dual anti-wound inductor and regulate the amount ofoutput electrical current, transforming electrical current parametersfrom a first coordinate space defined by a first electrical currentthrough a first winding of the dual anti-wound inductor and a secondelectrical current through a second winding of the dual anti-woundinductor to a second coordinate space defined by the magnetizingelectrical current and the output electrical current.
 14. The method ofclaim 9, further comprising implementing an observer configured toimprove accuracy of a measurement of a first electrical current througha first winding of the dual anti-wound inductor and a second electricalcurrent through a second winding of the dual anti-wound inductor byestimating the first electrical current when the first electricalcurrent is out of phase for measurement and by estimating the secondelectrical current when the second electrical current is out of phasefor measurement.
 15. The method of claim 9, further comprisingimplementing a current control loop for control of the power converter,and is further configured to measure, as feedback parameters to thecurrent control loop: a first average current of a first winding of thedual anti-wound inductor over the duration of a pulse-width modulationperiod of the first winding; and a second average current of a secondwinding of the dual anti-wound inductor over the duration of apulse-width modulation period of the second winding.
 16. The method ofclaim 9, wherein the first winding and the second winding areconstructed such that the coupling coefficient is greater thanapproximately 0.70.
 17. A system for controlling an electrical currentthrough a dual anti-wound inductor integral to a power converter, thedual anti-wound inductor having a first winding and a second windingconstructed such that its windings generate opposing magnetic fields inits magnetic core and constructed such that a coupling coefficientbetween the first winding and the second winding is less thanapproximately 0.95, the system comprising: an input for receiving areference input signal; and a current control subsystem configured to:minimize a magnitude of a magnetizing electrical current of the dualanti-wound inductor to prevent core saturation of the dual anti-woundinductor; and regulate an amount of output electrical current deliveredby the power converter to the load in accordance with the referenceinput signal.
 18. The system of claim 17, wherein the power converter isa boost converter.
 19. The system of claim 17, wherein the referenceinput signal is a reference voltage indicating a desired voltage levelto be driven to the load.
 20. The system of claim 17, wherein thecurrent control subsystem is further configured to control themagnetizing electrical current and the output electrical currentindependently from one another.
 21. The system of claim 17, wherein thecurrent control subsystem is further configured to, in order to minimizethe magnitude of the magnetizing electrical current to prevent coresaturation of the dual anti-wound inductor and regulate the amount ofoutput electrical current, transform electrical current parameters froma first coordinate space defined by a first electrical current through afirst winding of the dual anti-wound inductor and a second electricalcurrent through a second winding of the dual anti-wound inductor to asecond coordinate space defined by the magnetizing electrical currentand the output electrical current.
 22. The system of claim 17, whereinthe current control subsystem comprises an observer configured toimprove accuracy of a measurement of a first electrical current througha first winding of the dual anti-wound inductor and a second electricalcurrent through a second winding of the dual anti-wound inductor byestimating the first electrical current when the first electricalcurrent is out of phase for measurement and by estimating the secondelectrical current when the second electrical current is out of phasefor measurement.
 23. The system of claim 17, wherein the current controlsubsystem comprises a current control loop for control of the powerconverter, and is further configured to measure, as feedback parametersto the current control loop: a first average current of a first windingof the dual anti-wound inductor over the duration of a pulse-widthmodulation period of the first winding; and a second average current ofa second winding of the dual anti-wound inductor over the duration of apulse-width modulation period of the second winding.
 24. The system ofclaim 17, wherein the first winding and the second winding areconstructed such that the coupling coefficient is greater thanapproximately 0.70.